Pre-equalisation for UMTS base station

ABSTRACT

An arrangement ( 100 ) and method for RF filtering in a Node B of a UMTS TDD system by providing: a DAC converter ( 130 ) converting digital signals to analog signals; providing a narrow band analogue channel filter ( 150 ) filtering the analog signals; and providing a digital pre-equalizer FIR filter ( 120 ) coupled before the DAC ( 120 ) to filter the digital signals, the digital pre-equalizer filter means substantially correcting for non linear phase response ( 122 ) non-ideality and amplitude response non-ideality ( 124 ) in the analogue channel filter ( 150 ). This provides the following advantage(s): it enables 3GPP Node B co-location specifications to be met while providing both good transmit accuracy and acceptable ISI performance; and it allows filter center frequency to be field tuned in software, permitting a basic RF single-channel filter to used with its center frequency being field adjustable to a desired value centered on a UMTS channel.

RELATED APPLICATIONS

This application is a continuation of U.S. patent application Ser. No.10/531,152 entitled PRE-EQUALISATION FOR UMTS BASE STATION and filed onApr. 7, 2006, which is hereby incorporated herein by reference in itsentirety.

FIELD OF THE INVENTION

This invention relates to RF (Radio Frequency) filtering, andparticularly (though not exclusively) to such filtering in wirelesscommunication applications.

BACKGROUND OF THE INVENTION

Specifications (3GPP TS 25.105 v3.10.0, ‘BS Radio Transmission andReception (TDD)’, hereinafter referred to as [1]) by the 3GPP (3^(rd)Generation Partnership Project) set out the performance of TDD (TimeDivision Duplex) Node B (base station in a 3GPP system) equipment. Thesespecifications cover the ‘Adjacent Channel Leakage Ratio’ (AMA) for NodeBs specified for equipment that is co-sited with other TDD or FDD(Frequency Division Duplex) Node Bs operating on adjacent channels.

For co-siting purposes, stringent specifications on the transmitspectral purity of UMTS TDD Node Bs call for a single channel RF filterto be fitted after the power amplifier (PA). The specification of the RFfilter is also extremely stringent and a very high Q passive filter isrequired in order to achieve the required stop-band. By adopting an RFfilter with such a steep roll-off factor, it is unavoidable that thefilter has an effect on the transmit accuracy; in fact, the inclusion ofthis filter may cause the Node B to fail the transmit accuracyrequirement.

Passive RF equalization is not desirable due to the correspondingincrease in the complexity of the RF filter and the fact the amplitudeequalization is not possible without increasing the insertion lossacross the pass band. Analogue baseband equalization is not desirable asthe equalizer response needs to be matched to the RF filter to achieveoptimum transmit accuracy and tuning baseband filters to match the RFfilter, which in practice would have a significant impact on theproduction of the Node B.

A need therefore exists for digital pre-equalizer for RF filter whereinthe abovementioned disadvantage(s) may be alleviated.

STATEMENT OF INVENTION

In accordance with a first aspect of the present invention there isprovided a filter arrangement, for use in a wireless communicationtransmitter, as claimed in claim 1.

In accordance with a second aspect of the present invention there isprovided a method, for filtering in a wireless communicationtransmitter, as claimed in claim 15.

BRIEF DESCRIPTION OF THE DRAWINGS

One digital pre-equalizer, arrangement and method for RF filteringincorporating the present invention will now be described, by way ofexample only, with reference to the accompanying drawing(s), in which:

FIG. 1 shows a block-diagrammatic representation of an exampletransmitter architecture showing application of digitalpre-equalization;

FIG. 2 shows a graphical representation of the magnitude response of asingle channel RF filter;

FIG. 3 shows a graphical representation of group delay improvementfollowing introduction of a phase equalizing digital filter;

FIG. 4 shows a graphical representation of an ideal modulation mask andRF filter amplitude roll-off;

FIG. 5 shows a graphical representation of an example of amplitudeequalized RF filter;

FIG. 6 shows a block-diagrammatic representation of an exampleimplementation of the digital pre-equalizing FIR filter of FIG. 1.

DESCRIPTION OF PREFERRED EMBODIMENT(S)

The 3GPP specifications [1] referred to above cover the ‘AdjacentChannel Leakage Ratio’ (ACLR) for Node Bs specified for equipment thatis co-sited with other TDD or FDD Node Bs operating on adjacentchannels.

ACLR is a measure of the ratio between the signal power transmitted inthe desired channel of operation and the unwanted power transmitted inthe channels adjacent to the desired channel. In the referenced versionof the specifications, the adjacent channel power is specified as anabsolute limit of −80 dBm in a measurement bandwidth of 3.84 MHz.

This limitation is necessary to ensure that the transmission of a Node Bin channel ‘A’ does not cause unacceptable interference to another NodeB receiving in channel ‘B’ at the same time.

In most Node Bs, the power transmitted in the adjacent channel isdetermined by the linearity of the power amplifier used in the Node Btransmitter. With the PA technology available today, it is not possibleto achieve these levels of adjacent channel power at typical Node Btransmit powers levels. A brief example follows to illustrate thisproblem.

It is assumed that the adjacent channel transmissions are related to thelevel of the 3^(rd) Order intermodulation products generated within thepower amplifier. Taking an example Node B power amplifier with P1 dBequal to 44 dBm, and output IP3 of +63 dBm, the maximum wanted transmitpower consistent with a −80 dBm adjacent channel power will typically beof the order of +15 dBm (31 mW) for a CDMA (Code Division MultipleAccess) test signal. The power amplifier will dissipate close to 100 W,representing a DC to RF power conversion efficiency of 0.03%.

In order to achieve the specified ACLR, it is clear from the aboveanalysis that, for reasonable transmit powers, a narrow band RF filteris required.

Using the power amplifier in the previous example as a reference, thetypical ACLR expected at a transmit power of +34 dBm (2.5 W) will be ofthe order of 55 dB (or −21 dBm absolute). With this level of adjacentchannel power generated in the PA, the RF Filter is required to provideat least 60 dB of protection to the adjacent channel.

This is also an extremely difficult specification to meet and requiresthe use of very high Q dielectric resonators.

In the considered base station transmitter the RF filter mustimmediately follow the power amplifier, and as such the filter must berealised using analogue techniques. It is well known that analoguefilters generally exhibit non-constant group delay, although it ispossible to approximate constant group delay at the expense of asignificantly relaxed roll-off rate. However, for co-siting purposes, avery steep roll-off rate is essential, resulting in conflictingrequirements between pass-band group delay variation and rate of filterroll-off.

Non-constant group delay has a direct effect on the quality of thetransmitted signal, as the different frequency components within thesignal experience different delays as they pass through the filter. Theresult is that the RF filter introduces inter-symbol-interference (ISI)to the transmitted signal.

The technical specifications in [1] define ‘Error Vector Magnitude’(EVM) as a measure of transmit accuracy. The EVM is a ratio of the idealreceived signal compared to the actual received signal, expressed as apercentage. The reference signal is filtered twice by a ‘square-rootraised-cosine’ (RRC) filter, once in the transmitter and once in themeasuring receiver; therefore, provided that there are no signalimpairments, the received reference signal should be ISI free. It may benoted that the receiver timing is optimised to minimise the EVM.

Simulations of the EVM obtained with the single-channel RF filterpresent have shown that the EVM is typically 17%. These examples onlyconsider the EVM contribution of the RF Filter, the rest of the transmitline up to this point is not included in this calculation.

The 3GPP specifications [1] specify the maximum EVM to be 12.5%;therefore it is clear that, even though the presence of the RF filter isrequired by one part of the specification, it causes a failure in theEVM part.

Although RF analogue equalization is possible, this approach is notpreferred for a number of reasons such as increased size, cost,insertion loss and complexity. Also, the analogue equalization isoptimised for the centre frequency of the filter, and it will be seenbelow that it is instead beneficial to optimise the equalizationdepending on the exact channel centre required by the application.

In order to achieve a suitable analogue equalization performance inorder to obtain an acceptable EVM contribution from the RF filter, thepassive equalizer will, typically, be almost as complex as the actualfilter itself.

Another problem with the passive RF analogue equalization approach isthat it is not suitable for realising amplitude equalizations withoutincreasing the insertion loss across the whole band of the filter. Thisis because passive implementations can only create attenuation, notgain.

Having ruled out passive RF equalization, the designer is left with thepossibility of baseband equalization. This can be accomplished eitherwith passive or active analogue baseband filters or with a digitalfilter. As will be explained in greater detail below, in the preferredembodiment of the present invention, the digital filter solution is theonly suitable solution in an application where the equalization requiredwill need to be optimised for each individual filter and where it willnot be possible to tune analogue equalizers in a production environment.The digital equalizing filter can be calculated by a computer programdirectly from a measurement of the RF filter to be equalized, thushaving minimal impact on the production of the unit.

The channel centre frequencies in the UMTS radio interface are definedto be an integer multiple of 200 KHz, however the 5 MHz channelallocations are nominally defined between integer blocks of 5 MHz, e.g.,1900 MHz to 1905 MHz. Obviously, the true centre frequency is 1902.5 MHzin this example, which is not an integer multiple of 200 kHz. The exactchoice of the centre frequency is up to the operator or the licensingauthority. For the example channel allocation, two possible centrefrequencies are possible: 1902.4 MHz or 1902.6 MHz.

As RF filters are expensive and have long lead-time for supply, it isnot desirable to have two different filters for each 5 MHz block ofspectrum. It is far more preferable to keep one filter in stock with thecentre frequency at the true centre frequency. The consequence of doingthis is that this centred filter will further degrade the transmitaccuracy of signals centred on 1902.4 MHz and 1902.6 MHz.

As will be explained below, in a preferred embodiment of the presentinvention, the amplitude equalizing section of the digital pre-equalizercan be easily set up in manufacture to be optimised for the specificcentre frequency. Also, if required, the coefficients can be changedremotely when the unit is in the field via software control to enablethe unit to transmit on either of the channel centres.

Referring now to FIG. 1, a transmitter architecture 100 is designed foruse in Node B equipment 200 of a TDD UMTS system (not shown). The Node Bequipment is suitable for co-siting. It will be understood thatco-siting covers:

-   -   a single antenna shared between TDD and FDD base stations    -   a single antenna shared between TDD and TDD base stations    -   each base station having its own antenna, but multiple base        station antennas occupying the same tower at the same cell site.

As will be explained in greater detail below, the transmitterarchitecture 100 incorporates digital pre-equalization utilising thepresent invention. The transmit architecture 100 includes a transmitfilter section 110, a digital pre-equalizer section 120, adigital-to-analog converter (DAC) section 130, a transmitter section140, and a post-conversion RF single channel filter section 150. I(In-phase) and Q (Quadrature-phase) components of a modulated transmitsignal are applied to respective root-raised-cosine (RRC) filters 112and 114 in the transmit filter section 110; the RRC filters 112 and 114have real filter coefficients. The outputs of the RRC filters 112 and114 are applied to a series arrangement of first FIR (Finite ImpulseResponse) digital filter 122 and a second FIR digital filter 124; theFIR digital filters 122 and 124, which have complex filter coefficients,will be described in greater detail below. The I and Q outputs from thesecond FIR digital filter 124 are applied to respectiveanalog-to-digital filters 132 and 134. The outputs of the ADC converters132 and 134 are applied to a transmit up-converter 142 to produce asingle transmit output signal of upwardly-translated frequency. Theoutput of the transmit up-converter 142 is applied to an RFsingle-channel filter 152, to produce an accurate and highlyband-limited transmit output signal T.

The function of the digital pre-equalizer section 120 is to correct forthe non-ideal passband characteristics of the single-channel RF filter152. These non-ideal characteristics can be resolved into two separatefactors:

-   -   Non-constant group-delay, which is equivalent to a non-linear        phase vs. frequency response; group delay variation is a        consequence of designing the filter with a very steep transition        region using a reasonable number of sections, and    -   Premature roll-off in the pass band of the signal—a consequence        of the filter design and practical realisation, i.e., a        consequence of finite Q.

Each factor can be considered individually.

Group delay equalization is achieved by making use of the knowledge thata filter with symmetrical impulse response has the property of linearphase. Consequently, the FIR digital filter 122 is constructed toprovide group-delay equalization by filtering the signal with atime-reversed version of the impulse response of the RF filter 152. Theimpulse response is obtained by applying the inverse discrete Fouriertransform on the measured frequency response of the RF filter 152.

A suitable equalizer is obtained by truncating and quantising theimpulse response. All the necessary processing can be readily computedby a typical desktop computer. It will be understood that the exactsignal processing scheme applied in order to correct the phase responseof the filter is not critical, and a suitable signal processing schemewill be within the knowledge of a person of ordinary skill in the fieldof the invention.

FIG. 2 shows a graphical representation of the magnitude response of asingle-channel, narrow-band RF filter used in a 1.9 GHz UTRA Node B.

FIG. 3 shows the improvement in group delay by pre-filtering thetransmitted data signal with the digital FIR filter 122, the upper andlower lines indicating the group delay with and without thepre-filtering respectively.

Although the phase-equalizing filter 122 provides sufficient correctionfor the non-linear phase response of the RF filter, the resultingimprovement in transmit accuracy (Error Vector Magnitude) becomeslimited by the amplitude roll-off in the pass-band. Therefore, it isnecessary to introduce a correction for the amplitude response.

For this reason the second FIR filter 124 is used, which attempts acorrection for amplitude response without impacting the phase correctionproperties of the first FIR filter 122. This criterion implies that thesecond, amplitude-correcting filter 124 must be a symmetrical FIR filterand thus exhibit linear phase.

This second filter 124 can also be used to make additional corrections,by correcting for asymmetrical RF filter response around the desired RFchannel centre frequency, thus allowing one RF Filter to be optimisedfor centres offset from the true RF filter centre frequency by a smallamount.

In the present example a single RF filter, centred on say 1902.5 MHz,can be optimised separately for channel centre frequencies of 1902.4 MHzand 1902.6 MHz, thus reducing the number of alternative RF filtersolutions required.

Being digital, the equalizing filter 122 is programmable, providing theability to optimise the filter response to permit a Node B to operate oneither of these two frequencies in the field (e.g., via softwarecontrol), without the requirement to change the RF filter.

It should also be noted that the phase-equalizing filter 122 alsoemphasises the amplitude roll-off of the RF filter as the signal iseffectively filtered twice; therefore, the inclusion of the phaseequalizer 122 increases the need for the amplitude equalizer 124.

FIG. 4 compares the amplitude response of the ideal modulation mask(narrower shape) and the RF filter (wider shape). It may be noted that a100 KHz offset exists between the RF filter and the modulation,resulting in more attenuation on the low side of the modulation.

Even though the amplitude roll-off is small (typically the RF filter hasrolled off by 1 dB at the 3 dB points on the modulation mask) the effecton EVM is significant. Characterization of several RF filters has shownthat applying only phase equalization results in EVMs around 8% (reducedfrom 17% for the un-equalized filter). Applying amplitude equalizationcan improve the error vector to an acceptable level of approximately 3%.

To design the FIR filter 124, a ‘least-squares’ filter design program isused. Such a design program is readily available in commercial software,and need not be described in further detail. As the requiredequalization response is asymmetrical and most commercial FIR filterdesign tools produce real-valued symmetrical FIR structures, the filteris designed as a pass-band filter which is then down-converted to acomplex low-pass structure suitable for implementation in the transmitdigital processing.

FIG. 5 shows an example of the result obtained from the amplitudeequalizer, the lower and upper lines indicating respectively filterresponse after phase equalization only and after both amplitude andphase equalization. As can be seen by comparing the two lines, asignificant improvement in the pass-band flatness is achieved.

The overall filter is simply obtained by convolving the impulse responsefrom the phase equalizer and amplitude equalizer. The length of thefilter is optimised by selecting the N consecutive coefficients thatcontain the highest accumulated energy. The number of taps, N, requiredis a function of the required equalization accuracy.

It will be understood that in the digitally pre-equalized transmitterarchitecture 100 described above and shown shown in FIG. 1, because thedigital FIR filtering 120 equalises for errors in the analoguesingle-channel RF filter 152, the RF filter 152 may be deliberatelydesigned to roll-off in the pass-band of the desired signal in order toachieve a specified stop-band attenuation for a smaller, cheaper RFfilter implementation.

It will be understood that in the digitally pre-equalized transmitterarchitecture 100 described above and shown shown in FIG. 1, thesingle-channel RF filter 152 is the final component in the transmitarchitecture. In a TDD system it is possible to use the RF filter 152for both transmit and receive functions. Although the receive signalprocessing is not shown, the digital equalizer can also be used thiscontext.

Unlike the fixed definition transmit filter (root-raised-cosine withroll-off factor of 0.22 in the case of UMTS), the digital pre-equalizerneeds to be fully programmable; therefore its associated implementationcomplexity is high in terms of its gate-count. Therefore, in practice,steps need to be taken to reduce the number of gates required.

The number of gates required to build each FIR filter are related to thelength of the filter (i.e., number of coefficients) and the quantisationof both the data path and coefficient values. Simulations can be used todetermine the optimum filter length and coefficient quantisationrequired based on a sample of RF filters.

By way of an example the response of several filters was measured andthe appropriate equalizers were designed.

It was found that the impulse responses for all the filters tested wassimilar, therefore the size of the multipliers used to implement thisfilter could be optimised based on the magnitude of the expected valueof the coefficients.

FIG. 6 shows an implementation of the digital pre-equalizer section 120based on the above example. As shown, the FIR digital filter (theoverall filter, referred to above, obtained by convolving the impulseresponse from the phase equalizer 122 and amplitude equalizer 124)consists of 40 stages (of which six are shown). Each stage receives a10-bit real value Re{x(n)} and a 10-bit imaginary value (Im{x(n)}representing the I/Q input signal x(n) to be filtered, and multiplies(in one of the multipliers 160) this received pair of values by a pairof values representing the real part Re{h_(eq)} and the imaginary partIm{h_(eq)} of a respective filter coefficient. The imaginary parts ofthe filter coefficients 1-40 are 5-bit values; the real parts ofcoefficients 1-15 are 5-bit values, the real parts of coefficients 19-25are G-bit values, and the real parts of coefficients 26-40 are 7-bitvalues. Consequently, the multiplier outputs of the stages produce pairsof 15-bit values (stages 1-15), 16-bit values (stages 16-25), and 17-bitvalues (stages 26-40). The outputs of the multipliers 160 are combinedin summer 170, whose output of a pair of 22-bit values is applied to abit-select unit 180, which produces a pair of 10-bit output valuesrepresenting the filtered I/Q signal.

It will be understood that the exact number of bits used in the filteris not important, but that it is desirable for the number of bits usedin the filter to be optimised to reduce the complexity.

It will be appreciated that, in the application of the present example,the coefficients of the equalizer filter are complex rather thanreal-only, and that as a consequence the filter is more complex. Afilter with complex input data and real-only coefficients has toimplement the same filtering on both real and imaginary input data;hence two multiplications are required for each I and Q data pair. Ifthe coefficients are complex, then each multiplier has to be a fullcomplex multiplier which results in four complex multiplications and twosummations for each I and Q data pair. Therefore it is a significantbenefit in terms of the complexity of the filter if the number of bitsin the multipliers are minimised. In this application, the degrees ofasymmetry in the filter response are small; therefore the filtercoefficients can be optimised so that the largest coefficients are real.This allows the imaginary coefficients to be small and hence require afewer number of bits. Also, only one part of the impulse response of thefilter has coefficients with large magnitude; therefore the size of theprogrammable filter can be optimised to the general form expected forthe equalizer response. It may be noted that each RF filter may requirea slightly different impulse response, and the number of bits assignedto each section of the filter must take this variance into account.

It will thus be understood that there is an advantage to be had for afilter implementation which implements an asymmetrical amplitude andphase response by using a complex-coefficient filter and in which thecoefficients themselves have been phase rotated to ensure that thelargest coefficients are real; hence for relatively small amounts offilter asymmetry, the implementation complexity of the filter isminimised.

It will be understood that the digitally pre-equalized RF filteringscheme described above provides the following advantages:

it enables 3GPP Node B co-location specifications to be met whileproviding both good transmit accuracy and acceptable ISI performance;and

it allows filter centre frequency to be field-tuned in software,permitting a basic RF single-channel filter to used with its centrefrequency being field adjustable to a desired value centred on a UMTSchannel.

The invention claimed is:
 1. A filter arrangement for use in a wirelesscommunication transmitter having a transmit section including a poweramplifier, the arrangement comprising: a digital signals receivingsection for receiving digital signals to be transmitted; adigital-to-analog (DAC) section for converting the digital signals toanalog signals; an analog channel filter section for filtering theanalog signals after processing by the power amplifier; and a digitalpre-equaliser filter section coupled before the DAC section forfiltering the digital signals, the digital pre-equaliser filter sectionbeing adapted to filter the digital signals: first, with a time reversedversion of the impulse response of the analog channel filter section, tosubstantially correct for non-linear phase response in the analogchannel filter section; and second to substantially correct for linearamplitude error response in the analog channel filter section.
 2. Thefilter arrangement claim 1 further comprising an upconverter sectioncoupled between the DAC converter section and the analog channel filtersection for providing upward frequency translation.
 3. The filterarrangement of claim 1 wherein the digital preequaliser filter sectionis adapted to adjust to a desired value for the centre frequency of theanalog channel filter section.
 4. The filter arrangement of claim 1wherein the digital preequaliser filter section is programmable.
 5. Thefilter arrangement of claim 1 wherein the digital preequaliser filtersection has complex coefficients to provide asymmetric equalisation. 6.The filter arrangement of claim 1 wherein the arrangement is adapted foruse in a received signal path.
 7. The filter arrangement of claim 1wherein the wireless communication system is a UMTS wirelesscommunication system.
 8. Node B equipment comprising the filterarrangement of claim
 1. 9. A method for filtering in a wirelesscommunication transmitter, the method comprising: receiving digitalsignals to be transmitted by a communication transmitter having atransmit section with a power amplifier; providing a digital-to-analog(DAC) section converting the digital signals to analog signals;providing an analog channel filter section filtering the analog signalsafter processing by the power amplifier; and providing a digitalpre-equaliser filter section coupled before the DAC section to filterthe digital signals, the digital pre-equaliser filter being adapted tofilter the digital signals: first, with a time reversed version of theimpulse response of the analog channel filter section, to substantiallycorrect for non-linear phase response in the analog channel filtersection; and second to substantially correct for linear amplitude errorresponse in the analog filter section.
 10. The method of claim 9 furthercomprising providing an upconverter section coupled between the DACconverter section and the analog channel filter section to provideupward frequency translation.
 11. The method of claim 9 wherein thedigital pre-equaliser filter section adjusts to a desired value for thecentre frequency of the analog channel filter section.
 12. The method ofclaim 9 wherein the digital pre-equaliser filter section isprogrammable.
 13. The method of claim 9 wherein the digitalpre-equaliser filter section has complex coefficients to provideasymmetric equalisation.
 14. The method of claim 9 further comprisingusing the DAC section, the analog channel filter section and the digitalpre-equaliser filter section in a received signal path.
 15. The methodof claim 9 wherein the wireless communication system is a UMTS wirelesscommunication system.
 16. The method of claim 9 wherein the method isperformed in Node B equipment.
 17. The method of claim 9 wherein thestep of providing the digital pre-equaliser filter section includes:performing measurements of the analog channel filter section; andautomatically calculating on the basis of the measurements coefficientsof the digital pre-equaliser filter section.
 18. The method of claim 9wherein the step of providing the digital pre-equaliser filter sectionincludes: providing quantised filter coefficients of the digitalpreequaliser filter section based on the impulse response of the digitalpre-equaliser filter section.